Static converter switch with fast recovery freewheel diode

ABSTRACT

A static converter, including at least one pair of mutually co-acting switching transistors (TR1, TR2), of which the first switching transistor (TR1) is controlled with a frequency exceeding the control frequency of the second switching transistor (TR2). A diode (D1, D2) is connected between the emitter and collector of respective switching transistors. The noval feature of the invention resides in that a voltage source (N1, P) is activated when the first switching transistor (TR1) is energized to its conducting mode (I R ) and to supply a reverse recovery current (I rr ) through the diode (D2) of the second switching transistor in the reverse direction of the diode.

The invention relates to an arrangement in a static converter, includingat least one pair of mutually co-acting switching transistors, of whichthe first switching transistor is arranged to be controlled with afrequency exceeding the control frequency for the other switchingtransistor, and in which converter a diode is connected between theemitters and collectors of respective transistors.

An arrangement of this kind is often used for switching high poweroutputs when loading inductively, and such an arrangement will bediscussed more clearly hereinafter. The problem which arises, and whichis solved, or at least greatly reduced, by means of the inventionconcerns the high losses which occur because one of the switchingtransistors in said transistor pair is pulse-width modulated with afrequency of, for example, 1-8 kHz, while the other transistor of saidtransistor pair is controlled with a much lower frequency, for example afrequency of 50-60 Hz. This means that the first switching transistor isrepeatedly switched on and off while the other transistor is in anon-conducting state or mode. The problem of losses created hereby,which problem will be discussed in more detail hereinafter, is solved inaccordance with the invention, by arranging for a voltage source to beactivated when the first switching transistor is in its conducting mode,and to feed a reverse recovery current through the diode of said secondswitching transistor in the reverse direction of said diode.

The known prior art and the present invention will be described in moredetail with reference to the accompanying drawings, in which

FIG. 1 illustrates the principal construction and operational mode of aso-called Darlington circuit;

FIG. 2 illustrates an equivalent circuit for a Darlington circuit ofmonolithic construction;

FIG. 3 illustrates a simplified static converter which has three pairsof switching transistors for operating a three-phase motor with anarbitrarily selected frequency;

FIG. 4 illustrates a pair of switching transistors in the circuit shownin FIG. 3;

FIG. 5 illustrates a known circuit for eliminating the problem ofshort-circuiting;

FIG. 6 illustrates another known circuit;

FIG. 7 illustrates a circuit according to the invention which has avoltage source for providing a reverse recovery current for the diode ofthe second switching transistor;

FIG. 8 illustrates the phase current when the first transistor in thecircuit shown in FIG. 7 is brought to a non-conducting state or mode;

FIG. 9 illustrates the passage of the reverse recovery current throughthe diode of the second transistor when the first transistor has begunto conduct current during a subsequent control pulse.

FIG. 10 illustrates the state of the circuit when the reverse recoverycurrent ceases;

FIG. 11 illustrates the state of the circuit when the control pulse tothe first switching transistor has ceased and commutation of the phasecurrent takes place to the second switching diode;

FIG. 12 illustrates the state of the circuit immediately prior to thesecond switching transistor being switched off; and

FIG. 13 illustrates a simplified arrangement according to the inventionfor producing a reverse recovery current for the diode of the secondswitching transistor.

Although it has been assumed that each pair of transistors andassociated diodes are incorporated in a monolithic structure, it will beunderstood that discrete components can also be used.

The so-called Darlington circuit is well known for switching high poweroutputs. The principal construction and operating mode of such a circuitis illustrated in FIG. 1, and is characterized by high currentamplification, because the effective current-amplification factor,calculated from the base of the transistor T1 to the collector of thetransistor T2 is equal to the product of the current-amplificationfactors of both transistors. In its most usual form, a Darlingtoncircuit comprises a monolithic transistor, i.e. a transistor in whichall components are formed on one and the same silicon plate.

An equivalent transistor circuit is illustrated in FIG. 2, and it willbe seen from the Figure that there is formed in the monolithic structurea parasite diode D₁. Although this parasite diode is able to conduct thesame current as the transistor, it has a relatively long reverserecovery time t_(rr), in the order of 1-10 μs, with respect toswitching. The advantage with the parasite diode is that it can be usedas a so-called free-wheel diode. A free-wheel diode is required when theDarlington transistor is used for switching the current to an inductiveload. In order to function under these circumstances without seriouslosses occurring and without the provision of a current-limiting net,the modulation technique must be such that a transistor is withoutvoltage when it is energized to its conductive mode, i.e. a Darlingtontransistor must be without voltage when it is energized, which limitsthe usefulness of the transistor. A well known modulation technique ispulse-width modulation (PWM), which requires the transistor to beenergized to conductive mode with a full voltage between collector andemitter, which means, for example, that the free-wheel diode D2 willconduct when the Darlington transistor TR1 shown in FIG. 3 is energizedand brought to the conducting mode. FIG. 3 illustrates an a.c.rectifying bridge having Darlington transistors TR1-D1, TR2-D2, TR3-D3,TR4-D4, TR5-D5, and TR6-D6, each equivalent to the circuit illustratedin FIG. 2. In accordance with known techniques, the base electrodes ofthe transistors are supplied with control signals which determine thefrequency of the alternating voltage. In the illustrated case a controlvoltage of high frequency is supplied to the input A to the firsttransistor TR1, TR3 and TR5 respectively, while a low frequency controlvoltage is supplied to the second transistor TR2, TR4 and TR6respectively in each pair TR1, TR2 and TR3 respectively, i.e. the firsttransistor of each transistor pair will be switched on and off a numberof times during the time when the second transistor of each transistorpair is throttled. The a.c. rectifying bridge supplies a three-phasemotor M via phase conductors R, S and T. The limitation of theDarlington transistor resides in the fact that the free-wheel diode, asbefore mentioned, has a relatively long reverse recovery time, whichresults in excess current in the first transistor when energized. Theproblem of overcurrent, or excess current, occurs in the followingmanner. If one considers FIG. 4, which illustrates a part circuit of thea.c. rectifying bridge illustrated in FIG. 3, and if it is assumed thatthe motor current I_(R) passes for the moment through the free-wheeldiode D2, it will be found that when the Darlington transistor TR1begins to conduct current, and thus take over the motor current I_(R),D2 will not be blocked until the reverse recovery time t_(rr) haslapsed, for the aforementioned reason. During this period of time, thediode D2 will conduct current in the reverse direction (reverse recoverycurrent) and in principal a short circuit occurs via TR1 and D2 and alarge current will flow in this short circuiting circuit. Variousattempts have been made to eliminate this disadvantage. One suchattempted solution has been to connect an external series diode D7 and arapid free-wheel diode D8, as illustrated in FIG. 5, in order to isolatethe parasite diode, for example the parasite diode D1.

Another known solution is to limit the current growth in the circuitwith the aid of inductances, so that the current peak occurring when thecircuit is energized has reached a controlled and acceptable level whenco-acting free-wheel diode is extinguished. Such a circuit isillustrated in FIG. 6. In this circuit, there is connected between theemitter of the transistor TR1 and the collector of the transistor TR2 aferrite core which is wound with winding N1 and inductance L. Connectedacross the winding N1 is a zener diode Z1 and a diode D9. The circuitD9-Z1 empties the inductance of the energy stored during the conductingperiod. As will be seen from the following, relatively large losses areobtained in this circuit. Another known method of emptying theinductance and reducing losses is to arrange a winding N2 in series witha diode D10, as indicated in FIG. 6. A circuit D9-Z1 will still berequired, however, due to the leakage inductance between the windings,with resultant losses.

When the transistor TR2 conducts current, the current passes through theinductance. The inductance is emptied of energy after each conductingperiod, and the losses in the net D9-Z1 are;

    P=L·I.sup.2 ·f/2

where

I=mean current in the inductance

f=transistor control frequency.

With

I=15 A

L=60 μH

f=3000 Hz

there is obtained P=20 W, which requires large passive components inorder to restrict temperature rises.

The disadvantages with the known circuits for eliminating or reducingthe harmful reverse recovery current of the free-wheel diode-parasitediode can be eliminated in accordance with the invention by supplyingthe free-wheel diode with a reverse recovery current from a separatecurrent source. As will be made apparent hereinafter, such currentsupply can be effected with simple means. In accordance with theinvention, the current load on the transistor which is energized isgreatly reduced and because the occurring current peak and itsderivative can be increased there is obtained a reverse recovery time ofshorter duration in respect of the free-wheel diode in the monolithicstructure of the Darlington transistor, enabling the use of magneticcomponents which are smaller, and thus less expensive, than thecomponents used in an arrangement according to FIG. 6. This is becauseit is a given charge which is to be emptied from the free-wheel diode.It is true that the charge increases with the current derivative,although not more than an increase of the derivative from 4 to 75 A/μsresults in at least a halving of the reverse recovery time t_(rr). Apreferred embodiment of a circuit according to the invention isillustrated in FIG. 7. This circuit differs from the known circuitillustrated in FIG. 6 in that the inductance for limiting current growthis replaced with a transformer, which is coupled between the collectorand emitter of the transistor TR2, across one, or optionally more,series-connected diodes D11, enabling a feedback current (reverserecovery current I_(rr)) to be conducted. The transformer winding N1forms the aforementioned separate voltage source.

The functional mode of the circuit according to the invention isdescribed with reference to FIGS. 8, 9, 10 and 11.

FIGS. 8 and 9 illustrate commutation from D2 to TR1. It is first assumedthat the current I_(R) flows through the free-wheel diode D2 to the lineR, which constitutes one of the three phases of the motor M. When thetransistor TR1 is energized to its conducting mode, the voltage acrossthe free-wheel diode D2 of the transistor TR2 is only some volts and thewhole of the supply voltage U lies across the winding N2. Since N1 andN2 are transformer-coupled, the voltage U·N1/N2 will lie across thewinding N1 and its polarity is such that the voltage on diode D2 isreversed and the major part (N2/N2+N1) of the reverse recovery currentI_(rr) will thus flow in the circuit D11-N1-D2. This assumes that thetransformer is not saturated during the reverse recovery time t_(rr) ofthe diode D2.

From the equation U=N·d.0./dt there is obtained the smallest area forthe ferrite core for avoiding saturation during the reverse recoverytime;

    A≧U·t.sub.rr /(B·N2)

A=cross-sectional area of the ferrite core

t_(rr) =reverse recovery time of the diode

B=flux density upon saturation of the ferrite core

with some typical values of the parameters there is obtained*

    A≧300·1.5·10.sup.-6 /(0.4·25)=45×10.sup.-6 m.sup.2

This core may, for example, be a standard toroid core having a diameterof about 25 mm and a winding ratio N2=25N1=2. With a transistor of thetype MJ 10016, this will provide a recovery current I_(rr) ≈60 Ampere,of which only 5 Ampere will pass through transistor TR1. No current ispassed in the net D9-Z1, and thus no losses occur in said net duringthis phase.

FIGS. 10 and 11 illustrate commutation from TR1 to D2. It is assumedthat the phase current I_(R) flows from TR1 to the motor phase R, thelower part of the circuit illustrated in FIG. 10 being without current.When TR1 is throttled, the current I_(R) will commute to the free-wheeldiode D2 (FIG. 11). Because of the inductance in the transformer windingN2, the current is forced to pass to the phase conductor R through thecircuit Z1-D9. The voltage across this circuit Z1-D9 means that thetransformer is saturated after a time period of

    t.sub.m =A·B·n2/(U.sub.Z1 32 U.sub.D5)

where

A=cross-sectional area of the ferrite core (m²)

B=flux density upon saturation of the ferrite core (V·s/m²)

n2=the number of winding turns of the winding N2

U_(Z1) =the voltage drop in volts across Z1 (V)

U_(D9) =the voltage drop in volts across D9 (V)

When the ferrite core is saturated, the phase current will flow via thecoil or winding N2. During the saturation time T_(m) there is obtained apower loss in the circuit Z1-D9, which with previously assumed values isabout 7 watts. With previously assumed values, the saturation time T_(m)is about 4μ. A further, small power loss is obtained at that moment whenthe transistor TR2 is blocked and thus becomes non-conducting. Thispower loss is illustrated in FIG. 12. The phase current I_(R) then flowsthrough the coil or winding N2 and through TR2 from the phase R. Whenthe energy stored in the transformer is consumed in the circuit Z1-D9 isswitched off, there is obtained a power loss which with theaforementioned toroid core and with other conditions unchanged will beless than 1 watt. Thus, the total power loss will be about 8 watts,which shall be compared with a power loss of 20 watts in an arrangementaccording to FIG. 6.

FIG. 13 illustrates a simplified voltage source P in the form of abattery, which is coupled to D2 by means of a switch S and produces thereverse recovery current I_(rr), in order to rapidly discharge D2, asdescribed above. The switch S, which is shown in its closed position,may comprise, for example, a transistor or some other well known switchelement, and the control is effective synchronously with the control ofthe first transistor TR1, and suitably the same control signal issupplied to the switch S as that supplied to the transistor TR1, asindicated in FIG. 13. Another possibility of producing the reverserecovery current through the diode D2 is to discharge a capaciterthrough said diode.

I claim:
 1. An arrangement in a static converter, including at least onepair of mutually co-acting switching transistors (TR1, TR2), of whichthe first switching transistor (TR1) is arranged to be controlled with afrequency exceeding the control frequency of the second switchingtransistor (TR2), there being connected between the emitter andcollector of respective switching transistors a diode (D1, D2),characterized by a voltage source (N1, P) arranged to be activated whenthe first switching transistor (TR1) conducts current, and to supply areverse recovery current (I_(rr)) through the diode (D2) of said secondswitching transistor (TR2) in the reverse direction of said diode.
 2. Anarrangement according to claim 1, characterized in that the voltagesource comprises a winding (N1) on a transformer (N1, N2) to which thesupply voltage (U) of the switching transistor is supplied when thefirst switching transistor (TR1) is energized to its conducting mode;and in that the winding (N1) is arranged to reverse bias the diode (D2)of the second switching transistor (TR2).
 3. An arrangement according toclaim 1, characterized in that said voltage source (P) is arranged to beenergized synchronously with the energization of the first switchingtransistor (TR1) to its conducting mode; and to reverse bias the diode(D2) of the second switching transistor.
 4. An arrangement according toclaim 2, characterized in that the transformer (N1, N2) has two windingsconnected in series with one another, of which one winding (N2) isconnected between the emitter of the first switching transistor (TR1)and the collector of the second switching transistor (TR2), and thesecond winding (N1) is connected between the collector and emitter ofthe second switching transistor (TR2).